System and method for chip timing synchronization in an adaptive direct sequence CDMA communication system

ABSTRACT

In a CDMA communication system (100) capable of communicating between a receiver (10) and a transmitter (20)direct sequence spread spectrum communication signals (30), a system and method for synchronizing receiver chip timing and transmitter chip timing. Transmitter (10) transmits a training bit sequence (31) coded with a spreading chip sequence. The receiver (20) adaptively determines a representation of a despreading chip sequence using a tapped delay line equalizer (400). The representation of the despreading chip sequence is represented by tap coefficient potentials of the equalizer (400). Receiver chip timing offset is determined based on the tap coefficient potentials.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is related to U.S. patent application "A System andMethod for Bit Timing Synchronization in an Adapative Direct SequenceCDMA Communication System" by Lee et al, 08/073,226, filed Jun. 4, 1993,now U.S. Pat. No. 5,297,162, issued Mar. 22, 1994), and U.S. patentapplication "A Communication Method for an Adaptive Direct Sequence CDMACommunication System" By Lee et al., 08/091,879, filed Jun. 7, 1993, nowU.S. Pat. No. 5,353,300, issued Oct. 4, 1994, both filed concurrentlyherewith, and both assigned to Motorola, Inc.

1. Technical Field

This invention relates in general to the field of communication methodsand synchronization in data communication systems and more particularlyto a direct sequence code division multiple access (DS-CDMA)communication system.

2. Background

Code division multiple access (CDMA) communication systems are usedextensively in satellite communications with military and commercialapplications. These systems are also known as spread spectrumcommunication systems because the communicated information is spreadover a wide allocated frequency spectrum. In CDMA communication systemsthe frequency spectrum can be reused multiple times.

Because CDMA modulation techniques are inherently more susceptible tofading conditions present at the terrestrial and land mobileenvironments, their application has been limited to satellitecommunications. However, with recent advancements in communicationsignal processing, CDMA communication systems are becoming increasinglypopular in terrestrial land mobile communication environments as well.For example, recent developments have allowed CDMA systems to be used incellular telephone communications environments.

In general, there are two CDMA types of communication systems. One isknown as frequency hoping CDMA system where the wide allocated spectrumis divide into a substantial number of narrower frequency band andinformation signal is switched or "hoped" over these frequency bands inaccordance with a predetermined code. The other CDMA system is known asa direct sequence CDMA communication system (DS-CDMA) where the userinformation signals in the form of binary bits are spread over theallocated frequency spectrum by combining them with spreading codesknown as pseudorandom noise (PN) codes. The spreading code comprises apredetermined sequence of binary states known as chips. Thus, whencombined, each user information bit interval gets coded with a spreadingchip sequence. Conventionally, a DS-CDMA transmitter produces a directsequence spread spectrum (DS-SS) communication signal by multiplying theuser information bit sequences by the spreading chip sequence.

Once received at a receiving end, the DS-SS communication signal isdecoded by multiplying the received signal by a despreading chipsequence having corresponding characteristics to the spreading chipsequence. In conventional DS-CDMA communication system, the receiverknows of the spreading chip sequence prior to start of a communicationcall. Thereafter, the receiver decodes the DS-SS communication signalbased on the known spreading chip sequence.

It is well known that in the presence of many users CDMA receivers inaddition to receiving the desired signal also receive manymultiple-access interfering signals. In presence of multiple accessinterference, reliable communication may be achieved when interferingsignals are received at approximately the same power level. When, thereis a large disparity in received signal powers, non-zerocrosscorrelations among the signals gives rises to a phenomenon known asnear-far problem. In near-far situations, higher power interferingsignals significantly degrade reception and decoding of a lower powerdesired transmission.

One conventional approach to improving the near far problem uses a powercontrol scheme where the powers from the receivers are 5 fed back andtransmitter powers are controlled to substantially remove the powerdisparity. In another approach, PN codes are constructed such that theyprovide orthogonality between the user codes, thereby reducing mutualinterference. This allows for higher capacity and better linkperformance. With orthogonal PN codes crosscorrelation is zero over apredetermined time interval resulting in no interference between theorthogonal codes provided only that the code time frames are alignedwith each other.

In conventional CDMA communication systems the spreading chip sequenceis either assigned by a self controller or it is pre-stored within thereceiving unit. As such, during despreading and demodulation process,the receiver knows of the spreading chip sequence. A more recentapproach for a CDMA receiver proposes an adaptive despreading ordemodulating process. In an adaptive CDMA system, the receiver isenabled to suppress multiple access interference by an adaptiveequalization process. In such a system, a CDMA transmitter transmits atraining bit sequence which is coded with the spreading chip sequenceand the receiver adaptively determines, based on the training sequence,the despreading code using a tapped delay line equalizer. Adaptivedetermination of the despreading chip sequence and suppression ofmultiple access interference allows significant number of users tocommunicate with each other over an spread spectrum channel withoutrequiring central control infrastructure, and as such paving the way forinfrastructureless communication systems.

However, in adaptive CDMA communication, the determined despreading chipsequence is not time synchronized with the transmitter because ofcertain time delays within the communication path or simply because thereceiver does not know when bit and chip timing of the transmitterstags. Conventional methods of determining bit timing and chip timingoffsets between the transmitter and receiver comprise performingcon-elation routines involving complex mathematical processingoperations. These operations are time consuming and therefore delayestablishment of communication link between transmitter and receiver.Therefore, there exists a need for a faster synchronization method whichcould be achieved in significantly shorter period of time than isachievable by conventional methods.

SUMMARY OF THE INVENTION

Briefly, according to the invention, there is provided a CDMAcommunication system capable of communicating between a transmitter anda receiver direct sequence spread spectrum communication signalscomprising binary bit sequences coded with a spreading chip sequence.Receiver chip timing and transmitter chip timing are synchronized bytransmitting a training bit sequence coded with the spreading chipsequence. The receiver adaptively determines a representation of adespreading chip sequence based on the training bit sequence using atapped delay line equalizer. The tapped delay line equalizer providestap coefficient potentials which represent the despreading chipsequence. Thereafter, the receiver determines receiver's chip timingoffset based on the tap coefficient potentials.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram of a CDMA communication system according to presentinvention.

FIG. 2 is a timing diagram of a DS-SS communication signal according topresent invention.

FIG. 3 is a block diagram of a CDMA transmitter used in communicationsystem of FIG. 1.

FIG. 4 is a block diagram of a CDMA receiver used in the communicationsystem of FIG. 1.

FIG. 5 is a block diagram of a spreading equalizer transmitter used inthe receiver of FIG. 4.

FIG. 6 is an exemplary timing diagram of a transmitter bit intervalcoded with a spreading chip sequence.

FIGS. 7-10 are exemplary timing diagrams showing effects of various chiptiming offsets on output of a chip matched filter used in the receiverof FIG. 4.

FIG. 11 is a timing diagram of decoded DS-SS communication signal ofFIG. 2.

FIGS. 12-13 are exemplary timing diagrams of out put of a summer used inspreading equalizer of FIG. 5 showing effects of various bit timingoffsets.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

While the specification concludes with claims defining the features ofthe invention that are regarded as novel, it is believed that theinvention will be better understood from a consideration of thefollowing description in conjunction with the drawing figures, in whichlike reference numerals are carried forward.

Referring now to FIG. 1, a communication system 100 embodying theprinciples of the present invention is shown. The communication system100 includes a plurality of CDMA transmitters 10 and a plurality of CDMAreceivers 20 which communicate direct sequence spread spectrum (DS-SS)communication signals 30. The DS-SS communication signal 30 comprises aradio frequency communication signal modulated with binary bits codedwith spreading chip sequence. The communication system 100 is anadaptive CDMA communication system whereby the despreading chip sequenceis adaptively determined after the CDMA receivers 20 demodulates theDS-SS communication signal 30. As described later in detail, thereceiver includes a tapped delay line equalizer which adaptivelydetermines the despreading chip sequence during a training interval.Because the adaptive equalization is performed in presence of multipleaccess interfering signals, it adaptively produces the despreading chipsequence which suppresses the effects of the multiple accessinterference and decode the DS-SS communication signal 30. Once CDMAreceivers 20 determines the despreading chip sequence communicationbetween the CDMA transmitter 10 may be carried on based on thedetermined despreading chip sequence provided the bit timing and chiptiming are synchronized.

In the present invention, the adaptive equalization during training isperformed without bit timing or chip timing synchronization of thereceiver and the transmitter. This is because performing synchronizationof any kind in presence of interfering signals is close to impossible.Thus, a redundant training bit sequence is transmitted to circumvent theneed for synchronization while the despreading chip sequence is beingdetermined during training interval.

Referring now to FIG. 2, a timing diagram of the DS-SS communicationsignal 30 as transmitted by the transmitter 10 of FIG. 1 is shown. TheDS-SS communication signal 30 comprises string of bits which are codedwith a spreading chip sequence. The bits and the chips are binarysignals assuming one of two states of +1 and -1 represented by voltagepotentials of V₊₁ and V₋₁ respectively. The V₊₁ and V₋₁ potentials areof equal magnitude but opposite polarity. In this description it isassumed that V₊₁ has a positive polarity and the V₋₁ has a negativepolarity. At the start of the DS-SS signal 30, a training sequence 31 istransmitted which is used by the receiver 20 to adaptively determinedespreading chip sequence using a tapped delay line equalizer based onthe training bit sequence. In the preferred embodiment of the invention,the training bit sequence comprises a predetermined redundant bitsequence having a non-alternating and continuous bit states, such as asequence of consecutive +1 bit state. The training sequence 31 isfollowed by a transmitter bit timing sequence 33 which is used tosynchronize receiver and transmitter bit timing. The transmitter bittiming sequence 33 is predetermined bit sequence having characteristicswhich gives the receiver information relating to the transmitter bittiming. As described later in detail, the transmitter bit timingsequence 33 comprises an alternating bit sequence having alternating bitstates of both +1 and -1. Following the transmitter bit timing sequence33, a user information sequence 35 comprising user generated data istransmitted. The user generated data carries the actual data forcommunication of which the transmission was initiated. The usergenerated data may for example be coded voice or raw binary data.

Referring now to FIG. 3, a blocked diagram of the CDMA transmitter 10 isshown. The CDMA transmitter 10, includes a central controller and signalprocessor block 220, which controls the entire operation of thetransmitter 10 including signal processing necessary for modulating andgenerating the spreading chip sequence. The transmitter 10, includes atraining sequence block 201 which generates the predetermined trainingsequence. The transmitter 10 also includes a transmitter bit timingsequence generator block 203 which generates the transmitter bit timingsequence following the training sequence. Finally, a user informationsequence block 205 provides user information in form of binary bitsequences. The user information may be originated from a variety ofsources, such as from a voice coder which receives voice informationfrom a microphone or it may comprise raw data information generated froma computing device. A selector block 207 0 under the control of thecentral controller and processor block 220 provides for selecting one ofthe training, bit timing or user information sequences in proper orderand applies it to a multiplier 209. A spreading chip sequence generatorblock 211 generates the spreading chip sequence to be combined with thebit sequence to be transmitted to the receiver. Preferably, thegenerated spreading chip sequence comprise well-known gold PN codeshaving desirable crosscorrelation and auto-correlation properties. Thespreading chip sequence has a predetermined number of chips (n) forcoding each bit of the transmission sequences. The multiplier 209multiplies one of the transmission sequences by the spreading chipsequence and applies it to a modulator 213. Modulator 213 may comprise anumber of well known binary signal modulators, such as binary phaseshift keying (BPSK) or quadrature phase shift keying (QPSK) modulators.Output of the modulator 213 is applied to a power amplifier 215 whichamplifies the modulated signal and applies it to an antennae 217 fortransmission. It may be appreciated that the block 220 and some of theother blocks described in conjunction with transmitter 10 maybeimplemented utilizing one or more of well known digital signalprocessors, such as DSP 56000 series manufactured by Motorola Inc.

Referring now to FIG. 4, the block diagram of the CDMA receiver 20 isshown. The spread spectrum communication signal is received at theantennae 301 and is applied to a preselector filter 303 which providesthe initial receiver selectivity. The filtered signal is applied to awell known base band demodulator 305. The base band demodulator 305comprises a well-known demodulator that demodulates the communicationsignal in accordance with the modulation scheme used in the transmitter10 to provide a baseband signal 306. The base band signal 306 is appliedto a well-known chip matched filter block 307. The chip matched filtercomprises a well-known integrate-and-dump or a low pass filter blockwhere the received DS-SS communication signal 30 is sampled andintegrated at chip rate and the result is dumped at the end of each chipinterval. The output of the chip matched filter is applied to adespreading equalizer 400 which, based on the training sequenceadaptively determines a despreading chip sequence. As described later indetail, the despreading equalizer provides despreading chip sequence byadaptively equalizing the detected coded bits with an uncoded prestoredsignal corresponding to the training bit sequence. A signal processorand controller block 320 performs all necessary signal processingrequirements for the receiver 20. The equalizer 400 despreads the DS-SScommunication signal 30 and provides a decoded communication signal atits output (415). The decoded communication signal is applied to a userinterface block 313 which may comprise one of a number of user interfacedevices such as a speaker, a computing device, a data display or a faxor voice mail machine.

Referring now to FIG. 5, a block diagram of the despreading equalizer400 is shown. The equalizer 400 comprises an n-tap delay line equalizerwhere, as mentioned before, n is the number of chips per bit in thespreading chip sequence. The tap delay line consists of a bank of n-1serially coupled flip-flops 402 with their outputs coupled to acorresponding number of multipliers 404. The bank of serially coupledflip-flops 402 operate as a shift register sequentially shifting, at thechip rate, sampled outputs of the chip matched filter 307, i.e., (r₁-r_(n)) during each bit interval. At the end of each bit interval, themultipliers 404 multiply the flip-flop outputs with tap coefficients C₁-C_(n) provided by a tap coefficient generator block 407. A summer 405sums the outputs of the multipliers 404 to provide the summer output408. As such, the summer output 408 represents integration of themultiplier outputs over one bit interval. The summer's output 408 isapplied to a comparator 409 and a threshold decision block 410. Thethreshold decision block 410 comprises a threshold comparator whichafter training interval provides the detected bits of the user bitsequence. The threshold decision block 410 provides the equalizer output415. The threshold detector decision block 410 determines the decodedbit state by comparing the summer output 408 with a bit state thresholdlevel. It may be appreciated that the equalizer output 415 and thesummer output 408 are related by having a (1/n) ratio therebetween.

During training, the comparator 409 compares the summer's output 408with a pre-stored sequence as provided by a block 403. The pre-storedtraining sequences is a pre-determined signal representing uncodedtraining sequence. Therefore, the training sequence comprises a signalsimulating uncoded redundant consecutive and non-alternating trainingbits. The comparator 409 compares the pre-stored training sequence withthe summer output and provides an error signal 411 which is applied to atap coefficient generator block 407. The tap coefficient generatorblocks uses either the Least Means Square (LMS) or Recursive LeastSquare (RLS) algorithm to update tap coefficients C₁ -C_(n) once everybit interval in order to minimize the error signal 411. The despreadingequalizer 400 updates the tap coefficient C₁ -C_(n) until the errorsignal between the detected bit sequence and the pre-stored trainingsequence is minimized. Hence, equalizing the summer output 408 with theoutput of the pre-store training sequence. As a result of equalizing thetransmitted training bit sequence and the pre-stored sequence, the tapcoefficients C₁ -C_(n) become a representation of the despreading chipsequence which despread the DS-SS communication signal 30 and suppressmultiple-access interfering signals without prior knowledge of thespreading chip sequence. As such, the tap coefficients C₁ -C_(n)represent of the despreading chip sequence. These coefficients are usedto despread the DS-SS communication signal 30 after the traininginterval has terminated.

Operationally, upon commencement of a transmission the receiver receivesthe training sequence 31 of the DS-SS communication signal 30 of FIG. 2.As mentioned, the training sequence comprises a bit sequence comprisingnon-alternating bit sequence, such as a bit sequence having continuouscoded states of either +1 or -1. Commensurate with the trainingsequence, the pre-stored sequence also presents continuous uncodedstates of either +1 or -1 during the training interval. When received,the training sequence is sampled at chip rate via the chip matchedfilter 307. The output of chip matched filter is applied to the tappeddelay line equalizer 400 where through recursive iteration of updatingthe tap coefficients C₁ -C_(n) the pre-stored training bit sequence anddetected bit sequence are equalized. When equalized, the produced tapcoefficients C₁ -C_(n) result in decoding or despreading of the DS-SScommunication signal 30 and elimination of the multiple accessinterfering signals. As such, the equalizer 400 produces tapcoefficients C₁ -C_(n) which are a representation of the despreadingchip sequence. Accordingly, the DS-SS communication signal 30 is decodedby adaptively determining a representation of the despreading chipsequence based on the training bit sequence.

After the despreading chip sequence is determined the resulting tapcoefficients despread the received DS-SS communication signal while alsoeliminating the interfering signals. It may be appreciated that afterthe training interval the summer's output 408 at the end of eachreceiver bit interval represents integration of the decodedcommunication signal over that receiver bit interval. The integration,as herein described, constitutes summation of multiplication resultduring discrete chip intervals. Ideally, when the equalizing tapcoefficients (C₁ -C_(n)) are determined after training, theirmultiplication by the chip matched filter outputs (r₁ -r_(n)) despreadsor decodes the incoming DS-SS communication signal. Therefore, thesummer's output 408 after each receiver bit interval is equal to thenumber of chips (n) multiplied by the bit potential of the decodedcommunication signal bit, i.e. V₊₁, or V₋₁ depending on the detected bitstate, i.e., whether the detected bit comprises +1 or -1.

It may be appreciated that the tap delay line equalizer 400 could beimplemented within the digital signal processor 320 of the receiver 20.As such the digital signal processor includes despreading means,determination means, comparison means and any and all other meansnecessary for processing and controlling to effectuate the requiredfunctions of the present invention as outlined in this specification.Alternatively the equalizer 400 may be implemented utilizingconventional digital and logical discrete components as is well known inthe art.

Because of propagation delays and the fact that the receiver does nothave any information relating to the start of a transmission, a probablediscrepancy between the receiver and the transmitter timing may existafter completion of the training interval. This timing offset may existboth for chip timing and bit timing of the receiver. Therefore, thedespreading chip sequence as provided by the tap coefficients C₁ -C_(n)may have to be synchronized for proper despreading of the DS-SScommunication signal 30. In the adaptive CDMA communication system ofthe present invention, after the training interval, a chip timing offsetestimation is made during a chip timing interval. This is because, asdescribed hereinafter, the chip timing offset information could beextracted from the tap coefficients of the equalizer.

CHIP TIMING OFFSET

According to chip timing aspect of the present invention, the voltagepotential or the energy stored in the tap coefficients C₁ -C_(n)includes chip timing information provided that the effects of theinterfering multiple access signals are suppressed. As described before,the despreading chip sequence is represented by the tap coefficients C₁-C_(n) and potentials thereof. In the communication system 100, theinterfering signals are eliminated after the training interval and upondetermination of the despreading chip sequence. Therefore, chip timingoffset determination is commenced following the training interval toalign receiver and transmitter chip timing. The chip timing offsetdetermination process of the present invention could take place duringone or more bit intervals after the final tap coefficients aredetermined.

Referring now to FIG. 6, an exemplary chip sequence during one bitinterval is shown. The chip sequence comprises n chips which assume oneof two states +1 and -1. Because the outputs of the chip matched filter305 (r₁ -r_(n)) when sampled by the receiver contain informationrelating to the receiver and the transmitter chip timing offset andbecause the voltage potentials representing the tap coefficients aredirectly proportional to the energy of the received chips at the end ofreceiver chip intervals, the tap coefficient potentials are processedfor determining the timing offset. Due to binary nature of the chipsequence, the ratio of the maximum potential of the outputs of the chipmatched filter 307 to the minimum output potentials relates to the chiptiming offset. According to the invention, the tap coefficientspotentials can be divided into two sets: one having maximum and anotherhaving minimum potentials. A first set of coefficient potentialscorresponds to those having maximum potentials (V_(max)) and a secondset of coefficient potentials corresponding to those having minimumpotentials (V_(min)). It has been determined that the voltage potentialof tap coefficients in second set change with respect to the tapcoefficients in the second set by a factor of (1-2 a), where arepresents the chip timing offset in terms of one chip interval. As suchthe following relationship exits between the chip timing offset and thetap coefficient voltage potentials:

    a=1/2*(1-|V.sub.min |/|V.sub.max |)Eq. (1)

Therefore, by examining the tap coefficient potentials relating to eachset the chip timing offset may be determined. It should be noted that inequation 1 the maximum potentials and the minimum potentials areexpressed in terms of absolute values. Therefore, their polarity isirrelevant for determination of chip timing offset.

To illustrate the above concept, a number of exemplary situations wherethe receiver bit timing off set is equal to zero, 1/2 chip interval,+1/4 chip interval and -1/4 of chip interval will be examined.

FIG. 7 shows the output of the chip matched filter 307 as it samples atchip rate, integrates during the chip interval, and dumps at the end ofthe chip interval when the transmitter chip timing and receiver chiptiming are synchronized, that is, chip timing offset=0. As shown, theoutput of the chip matched filter at the end of each intervals 701-707has one of two equal but opposite potentials V₊₁ and V₋₁. The potentialscorrespond respectively to either of the +1 or -1 potential of the chipstate. Because the sampled values r₁ -r_(n) are directly proportional tothe tap coefficients C₁ -C_(n), the absolute value of the first set ofcoefficient potentials, i.e., |V_(max) |, is equal to the absolute valueof the second set of potentials, i.e., |V_(min) |. Therefore, the ratioof |V_(min) |/|V_(max) | is equal to 1 resulting in a timing offsetdetermination of a=0 according to equation 1. As such, the timing offsetmay be determined by processing the tap coefficient potentials at theend of each bit interval. It should be noted that V_(max) (or V_(min)for that matter) as referred herein could be considered as correspondingto either one of V₊₁ or V₋₁ since the absolute values of the V_(max) orV_(min) are of significance equation (1).

Referring to FIG. 8, a receiver chip timing offset of 1/2 chip isassumed. That is, the chip interval 701 is half a chip off from the chipinterval 801. As shown, the output of chip matched filter at the end oftime interval 801 reaches V₊₁. Then at the end of time interval 802 thechip matched filter output reaches a zero potential. At the end of chipinterval 803, the output reaches V₋₁. Again, at the end of chipintervals 804-806, the outputs are at zero. And finally at the end ofinterval 807 the output reaches V₊₁. Accordingly, the first set ofcoefficient would have a potential V_(max) which is equal to V₊₁ (i.e.,V_(max) =V₊₁ (or V₋₁)) and the second set of coefficients V_(min) wouldhave a potential equal to zero (i.e., V_(min) =0). Therefore, fromequation (1) a chip timing offset of a=1/2 chip interval would result.

Referring to FIG. 9, a receiver chip timing offset of +1/4 is assumed.The positive sign of the chip timing offset signifies that thetransmitter chip timing leads the receiver chip timing. That is, thetransmitter chip timing reference starts prior to the receiver chiptiming reference. Following the above analysis V_(max) is equal to V₊ 1(or V₋₁) and V_(min) is equal to 1/2 of V₊₁. As such, the equation (1)yields a timing offset a=1/4 chip timing.

The timing offset determined based on Equation (1), however, does notprovide information relating to whether the timing offset is positive,or negative. The sign of the timing offset indicates whether thereceiver chip timing is leading or trailing the transmitter timingoffset. According to the invention, the sign of information can bedetermined by examining the polarity and magnitude of successive tapcoefficient potentials during one bit interval or two successive bitintervals. Therefore, once the absolute value of the timing offset a isdetermined further processing of the tap value coefficients results indetermination of the timing offset sign.

It may be appreciated that when the timing offset is equal to 1/2 chipinterval the sign of the offset becomes irrelevant since the receiverchip timing could be adjusted by one half chip interval in positive ornegative direction resulting in synchronization with transmitter chiptiming. Furthermore, positive chip timing offset of greater than 1/2chip timing offset could be expressed in terms of a negativecomplementary offset. For example, a positive 3/4 timing offset could beexpressed as a -1/4 timing offset and so on. Therefore, the timingoffset a would be a value within the range of zero to 1/2 with theoffset timing sign signifying the leading or trailing status of thereceive chip timing offset.

Referring to FIG. 10, a receiver chip timing offset of -1/4 is shown. Inorder to better understand the process by which sign of the timingoffset may be determined, the -1/4 timing offset of FIG. 10 will becompared with the +1/4 timing offset condition of FIG. 9. As can be seenin FIG. 9, during consecutive intervals 901,902 and 903, when there is atransition from a V₊₁ to a V₋₁ (chip transition of positive potential tonegative potential), the outputs of the chip matched consist of V₊₁,1/2V₊₁, and V₋₁. Due to the fact that the timing offset is positive inFIG. 9, after completion of a positive to negative chip transitionoccurring during intervals 601 to 602 (shown in FIG. 6), the output ofthe chip matched filter reaches a positive polarity, i.e., 1/2V₊₁, atthe end of interval 902. Conversely, in FIG. 10, because of negativetiming offset, the output of the chip matched filter after completion ofthe same positive to negative chip transition would reach a negativepolarity, i.e., 1/2V₋₁, at the end of interval 103. Therefore, the signof the timing offset could be determined based on the polarity of atleast one of the tap coefficient potentials after one or more chiptransitions. It may be appreciated that the same type of analysis isapplicable to a negative to positive chip transition as well as otherchip sequence arrangements.

The tap coefficient potential processing needed for determination 0 ofthe chip timing offset a and its sign could all be accomplished byappropriately programming the digital signal processor 320 utilizingwell known signal processing techniques. As such the signal processor320 includes means for determining chip timing offset based on the tapcoefficient potentials as well as the means for determining sign of thechip timing offset based on the polarity of at least one of the tapcoefficients after a chip transition.

Upon determination of the chip timing offset and sign thereof, thereceiver chip timing could be adjusted to synchronize it with thetransmitter chip timing. It should be noted that because of existence ofmultiple access interference the chip timing offset determinationaccording to the present invention produces an estimate and not sheprecise chip timing offset. Therefore, there may still be a need toperform some minor correlation routines to complete chipsynchronization. However, the amount of time needed to perform suchroutines is minimal. Once the chip timing synchronization is performed,the receiver 20 commences a bit timing synchronization process during abit timing interval.

BIT TIMING OFFSET

Because of redundancy of the training sequence no synchronization isnecessary during training interval. Assuming that the chip timing issynchronized, bit timing offset between the receiver and the transmitterduring the training interval causes the resulting tap coefficients C₁-C_(n), which represents the despreading chip sequence, to be cyclicallyshifted by a corresponding number of chips. Therefore, the receiver bittiming offset may be expressed in terms of chip numbers. When the DS-SScommunication signal 30 is despreaded, after the determination of tapcoefficient C₁ -C_(n) the resulting decoded DS-SS communication signalincludes bit timing information which may be extracted in conjunctionwith the transmitter bit timing sequence 33 of FIG. 2.

After the training interval, the summer output 408 provides arepresentation of the decoded DS-SS communication signal 30. Therefore,the summer output 408 is processed to determine the bit timing offset.

It has been determined that the following relation ship exists betweenthe bit timing offset and the summer output:

    y.sub.t =b.sub.t-1 (n-m)+b.sub.t (m)                       Equation (2)

Where y_(t) is the summer output at time t, b_(t-1) and b_(t) aredecoded bits at times t-1 and t, m is bit timing offset in terms ofnumber of chips, and n is number of chips per bit. When b_(t-1) andb_(t) are consecutive non-alternating bits, then y_(t) is equal to theirbit state, i.e., either +1 or -1. When b_(t-1) and b_(t) are alternatingbits then y_(t) =+(n-2m) if b_(t-1) =+1 and b_(t) =-1, and y_(t)=-(n-2m) if b_(t-1) =-1 and b_(t) =+1. Accordingly, the bit timingoffset information may be extracted by processing the summer output 408after an alternating transition from one bit state to another bit statehas occurred.

Referring to FIG. 11, a decoded communication signal upon completion ofthe training interval and after determination of the despreadingsequence is shown. As shown, the transmitter bit timing sequence 33follows the training sequence. The transmitter bit timing sequence whenreceived, provides the receiver 20 with the capability of detectingstart of the transmitter bit interval. The transmitter bit timingsequence 33 comprises a sequence of alternating bit sequence with atleast two consecutive bits having alternating states such that the stateof one bit changes from one interval to the succeeding interval. Inother words, a transition from +1 to -1, or vice versa, would existbetween two consecutive bits from a first bit interval to the subsequentsecond bit interval. The transitions occurring over the transmitter bittiming sequence are critical because they are indicative of transmitterbit timing which is used in the receiver to determine the bit timingoffset according to equation 2. As shown, an exemplary transmitter bittiming sequence may consist of the sequential bit states of +1, +1, -1,-1, +1, +1, -1, -1 occurring respectively in transmitter bit intervals111, 113, 115, 117. It may be appreciated that the transmitter bittiming sequence may be of other variety of sequences, such asalternating bit sequence of +1, +1, -1, 1, +1, -1 as long as thesequence consists of transitions conveying transmitter bit timinginformation.

In the preferred embodiment, the receiver bit timing offset isdetermined after the training interval by integrating, over a firstreceiver bit interval, non-alternating bits of the decoded communicationsignal to produce a first result, and integrating over, a secondreceiver bit interval, alternating consecutive bits of the decodedcommunication signal to produce a second result. Thereafter, the firstresult is compared with the second result to determine the bit timingoffset. The bit timing offset is determined by determining half thedifference between the first result and the second result. It should benoted that the first result may be a prestored constant valuerepresenting the result of integration over non-alternating bits.

The above concept may be better understood by referring to FIG. 12,where the summer output in a situation where the bit timing offset is -mchips is shown. The negative sign of the timing offset indicates thatthe transmitter bit interval occurs before the receiver bit interval.The normalized output of the summer at the end of the first receiver bitintervals which occurs during the consecutive non-alternating bit statesof +1, corresponding transmitter bit intervals 111 and 113, is equal ton, i.e. first result=n. The normalized summer output 408 at the end ofthe second receiver bit interval after integration during thealternating bit transition from +1 to -1 occurring on the transmitterbit interval 115 is equal to -(n-2m), i.e., second result is n-2m.Therefore, by determining half the difference between absolute values ofthe first result and the second result the absolute value of the bittiming offset m is determined.

The absolute value of the chip timing offset as determined above doesnot indicate sign of the receiver bit timing offset. In order to betterunderstand the process by which the sign of bit timing offset isdetermined, an exemplary situation where the receiver bit timing isequal to +m chips as shown in FIG. 13, is compared to the situation ofFIG. 12 where the receiver bit timing is -m chips. In FIG. 12, during apositive to negative transition (occurring during transmitter bitintervals 113 to 115), a negative bit timing offset produces a firstresult which has a positive polarity and a second result which has anegative polarity after the transition. Whereas, in FIG. 13, during thesame positive to negative transition, a positive bit timing offsetproduces a positive polarity first result and second result. In anotherexample, referring to FIG. 12th negative to positive transition from thetransmitter bit interval 117 to 119, produces a negative polarity firstresult (the polarity of the summer output as a result of integration oftwo consecutive -1s of intervals 115 and 117), and a positive secondresult when the bit timing offset sign is negative. In FIG. 13, duringthe same negative to positive transition, a positive bit timing offsetsign produces negative first and second results. As such, it may beappreciated that the sign of the bit timing offset may be determined bydetermining the type of transition, i.e. positive to negative or viceversa and comparing the polarities of the first result and the secondresult. Accordingly, of the bit timing offset is determined by comparingresults of integration produced during consecutive non-alternating bitswith that obtained during alternating bits.

Upon determination of the bit timing offset and sign thereof, thereceiver bit timing could be adjusted to synchronize it with thetransmitter bit timing. It should be noted that because of existence ofmultiple access interference the bit timing offset determinationaccording to the present invention produces an estimate and not theprecise bit timing offset. Therefore, there may still be a need toperform some minor correlation routines to complete bit synchronization.However, the amount of time needed to perform such routines is minimal.Once the bit timing synchronization is completed, the receiver 20commences to decode user information sequence.

As discussed above, the chip timing synchronization in the adaptive CDMAcommunication system 100 is achieved mainly based on informationinherent in the tap coefficients C₁ -C_(n) without performing complexcorrelation routines. Simple processing of the tap coefficientpotentials and derivation of the chip timing offset based thereonrequires substantially less time than the more time consumingcorrelation processing proposed in the prior art. As a result, asubstantially quicker and more efficient communication link between thereceiver and the transmitter is established.

While the preferred embodiments of the invention have been illustratedand described, it will be clear that the invention is not so limited.Numerous modifications, changes, variations, substitutions andequivalents will occur to those skilled in the art without departingfrom the spirit and scope of the present invention as defined by theappended claims.

What is claimed is:
 1. In a direct sequence CDMA spread spectrumcommunication system, wherein spread spectrum communication signalscomprising binary bits coded with spreading chip sequences arecommunicated between a transmitter and a receiver; a chipsynchronization method for synchronizing receiver chip timing withtransmitter chip timing comprising the steps of:a) transmitting a directsequence spread spectrum communication signal including a redundanttraining bit sequence; b) receiving said predetermined redundanttraining bit sequence; c) adaptively determining a representation of adesk, reading chip sequence based on said redundant training bitsequence using a tapped delay line equalizer; said despreading chipsequence being represented by tap coefficient potentials of saidequalizer; and d) determining receiver chip timing offset based on saidtap coefficient potentials.
 2. The method of claim 1, wherein said stepof determining the receiver chip timing offset includes the step ofdetermining a ratio of minimum tap coefficient potential to maximum tapcoefficient potential.
 3. The method of claim 1, further including thestep of determining the sign of the chip receiver timing offset based onthe polarity of at least one of the tap coefficients after a chiptransition.
 4. In a CDMA receiver capable of receiving spread spectrumcommunication signals comprising binary bits coded with spreading chipsequences form a transmitter; a bit synchronization method forsynchronizing receiver chip timing with transmitter chip timingcomprising the steps offa) receiving a redundant training bit sequence;b) adaptively determining a representation of a despreading chipsequence based on said redundant training bit sequence using a tappeddelay line equalizer, said despreading chip sequence being representedby tap coefficient potentials of said equalizer; c) determining receiverchip timing offset based on said tap coefficient potentials.
 5. Themethod of claim 4, wherein said step of determining the receiver chiptiming offset includes the step of determining a ratio of minimumcoefficient potential to maximum coefficient potential.
 6. The method ofclaim 4, further including the step of determining the sign of the chiptiming offset based on the polarity of at least one of the tapcoefficients after a chip transition.
 7. A CDMA communication system forcommunicating direct sequence spread spectrum communication signalscomprising:a transmitter for transmitting a direct sequence spreadspectrum communication signal including a redundant training bitsequence; a receiver for receiving said direct sequence spread spectrumcommunication signal including: a tapped delay line equalizer foradaptively providing tap coefficient potentials representing adespreading chip sequence based on said redundant training bit sequence;determination means for determining receiver chip timing offset based onsaid tap coefficient potentials.
 8. The CDMA communication system ofclaim 7, wherein said determination means includes means for determininga ratio of minimum tap coefficient potential to maximum tap coefficientpotential.
 9. The CDMA communication system of claim 7, furtherincluding means for determining the sign of the chip timing offset basedon the polarity of at least one of the tap coefficients after a chiptransition.
 10. A direct sequence CDMA receiver, comprising:means forreceiving a direct sequence spread spectrum communication signalincluding a redundant training bit sequence; a tapped delay lineequalizer for adaptively providing tap coefficient potentialsrepresenting a despreading chip sequence based on said redundanttraining bit sequence; determination means for determining receiver chiptiming offset based on said tap coefficient potentials.
 11. The receiverof claim 10, wherein said determination means includes means fordetermining a ratio of minimum tap coefficient potential to maximum tapcoefficient potential.
 12. The receiver of claim 10, further includingmeans for determining the sign of the chip timing offset based on thepolarity of at least one of the tap coefficients after a chiptransition.